When you have two nearly identical sine wave signals and you want to compare them, one technique is to plot one against the other, creating what is known as a Lissajous curve.

Lissajous curves make nice images, and even nicer videos because of the phase shifts.

So let’s take two signal generators and try it out, eh?

On the X-axis, I’m going to plot a 10 MHz sine wave from the new AWG, described on this weblog a few days ago. The frequency accuracy and stability of its output signal is within 1 or 2 ppm, according to the TG2511 specs.

On the Y-axis, let’s connect a second 10 MHz sine wave from a cheap DDS, also described on this weblog a few months back. This has a simple crystal so I’d expect 50 to 100 ppm frequency accuracy, i.e. within ≤ 1000 Hz.

When you connect these to an oscilloscope and put it in X-Y mode, you get pictures like these:

The more the two signals are in phase, the more the result will look like a straight line, slanted at 45° (from the bottom left to the top right). When exactly 180° out of phase, it will show a straight line from top left to bottom right. Everything in between create ovals, and when the signals are 90° out of phase (either lagging or leading), the result is a perfect circle.

So the big thing about Lissajous curves is that they let you compare the relative phase of two sine waves.

In practice, signals from different sources will tend to change phase over time, i.e. “drift” as one sine wave is slightly slower or faster than the other. This creates a way to precisely compare two frequency generators: measure how long it takes for the phase to go from 0° to 180° (or 360°, which is 0° again), and you get an idea how long it takes for one signal to catch up (or lag) one full sine wave over the other. Trouble with this approach is that sometimes these cycles are too fast to see, let alone time manually.

With an adjustable frequency source, there’s also another way: adjust a known frequency until the shape stays the same, and you’ll have “measured” the frequency of the other signal in terms of the adjusted one since they must now be equal. It’s very much like tuning a musical instrument by ear and adjusting for a “zero beat”.

That’s what I did, and I ended up with the following result for this test setup:

IOW, the cheap DDS is running 0.03% slow – i.e. about 300 ppm! And it’s not even very stable, because very soon the DDS starts drifting again: an indication that it’s not holding its frequency really accurately either. This is not really surprising for such a low-cost unit off eBay – it’s still a useful signal source: lots of useful experiments and measurements can be done with such a fairly decent 0.03 % accuracy level, after all.

Ok, this concludes my first exploration into signal-processing – enough signal theory for now!

Triggered by the recent signal generator checks, and those FM radio stations creeping into the signal yesterday, I wanted to do another test to see how and when this happens, using a series of scope FFT snapshots.

Here’s a 50 Ω coax cable of 2m length hooked up, with 50 Ω termination on both sides but no signal. The scale is 5 dBm per division, and I’ve zoomed into this very low range with a baseline of about -105 dB. The following 200 MHz wide FFT measurements were all done with the scope input set to max sensitivity, i.e. 1 mV/div:

Note the slight FM radio station RF signal pickup, even with a fully terminated coax cable!

Same thing, but disconnected on one end, i.e. only one 50 Ω terminator inside the scope:

I don’t know why there’s a peak at 25 MHz – the signal generator was completely powered down.

Here’s the spectrum with no coax at all, i.e. nothing connected to the scope, but its 50 Ω shunt still enabled:

When also adding an external 50 Ω terminator, the lower frequencies drop ever so slightly further:

And here’s what happens when the 2m 50 Ω coax cable is attached back on, without the 50 Ω termination:

As you can see, the coax cable now acts as antenna, picking up a few more signals at 38.45 MHz and 46.38 MHz. And FM reception shooting up to 20 dB above the noise floor. Even though it’s shielded!

The slight drop in noise across the screen from 0 to 200 MHz is probably nothing more or less than the scope’s bandwidth: a 200 MHz scope is specified as having a 3 dB drop at 200 MHz, which fits amazingly well with what all the above screen shots are showing.

These tests confirm the superb signal processing specs of the Hameg oscilloscope front end: a -105 dB noise floor @ 1 mV/div maximum sensitivity. For even lower noise levels (and a higher frequency range, as would be needed for 868 MHz and 2.4 GHz RF measurements), probably only a “real” spectrum analyzer will do better.

Whee… with a multi-meter probe wire attached as antenna, I can easily pick up all major AM radio stations:

Tomorrow, I’ll close off with one more post about signal processing: accurate frequency measurements.

Update – As with yesterday’s post, these FFT’s were produced with the Rectangle window function. As a bonus, here’s the frequency spectrum produced by the noise generator in my new AWG:

Down by 30 dB at 50 MHz, but a pretty good source of white noise at lower frequencies. The AWG can add an adjustable amount of this noise to the generated waveforms – can be useful to see how well a filter, demodulator, or other detector behaves, for example.

As shown in yesterday’s post, once you’re looking at signals in the high MHz range, it’s easy to make mistakes. Looking at that screen shot again, you can see a whole bunch of 90..100 MHz spikes:

These are in fact local FM radio stations – being picked up by my clunky scope probe hookup to the AWG. In other words it’s an antenna: radiated RF signals are accidentally being received by the probe and mixed in with the conducted 25 MHz signal. The way to get rid of these is to use “shielding” to keep those radio waves out.

Here’s the same signal, using a 50 Ω coax cable all the way between signal generator and oscilloscope:

No more weird stuff, just the 25 MHz multiples – indicating that there’s a slight distortion in the generated 25 MHz sine wave. This is normal for any signal generated by a direct digital synthesizer, because the waveform is created through through a digital-to-analog converter, fed at 125 million samples per second in this case. IOW, it’s an approximated sine wave (20 datapoints per sine wave @ 14-bit resolution).

So the big change is that the FM radio stations are gone, and that the signal’s noise level is now in fact a few dB cleaner than before – as you can see from the slightly lower tail towards 200 MHz.

I did have to use a small trick to make these graphs comparable: the second one has the scope’s 50 Ω internal terminator enabled, so that the path from signal source to signal destination is now done by the book: a 50 Ω source (in the AWG), feeding a 50 Ω coax cable, terminated by 50 Ω at the destination (in the scope). This does “attenuate” (i.e. reduce) the signal level by half, so I had to raise the baseline by 3 dB on the second FFT screen shot to make the height of the 25 MHz peak identical in both screens.

One other minor difference is that the second graph is smoothed over 256 samples i.s.o. 64 – cleaning up the resulting line slightly more.

So you see… it’s possible to do RF-type stuff without understanding all the details – which I certainly don’t, yet – and get decent results. The 25 MHz wave coming into the scope is very clean: the first harmonic is some 50 dB below the signal itself, which means that the first harmonic has 100,000 times less energy than the main signal.

Tomorrow, another post about this topic: cables, termination, and noise…

Update – As John Beale pointed out in the comments below, the FFT baseline is caused by the choice of FFT windowing function. Here’s the 50 Ω coax example again, using a Hanning window:

Much better for comparing relative dB differences between peaks.

(Tomorrow’s post will also use the default Rectangle window, sorry about that…)

Earlier this week, I described how a fixed frequency can be used to stabilize others.

Well… as part of my continuing drive to set up a more complete workbench here at JeeLabs, I’ve decided to get another piece of equipment which relies on this mechanism, called an “Arbitrary Waveform Generator” (AWG) or “function generator” or “signal generator” – three names for essentially the same instrument, as far as I can tell.

An AWG produces a repetitive electrical signal, such as a sine wave or a square wave. Very roughly speaking, you can think of sine waves as “pure analog” and square waves as “pure digital” frequencies.

The unit I picked is a fairly advanced one, the TG2511 from TTi (Thurlby Thandar Instruments), in the UK:

(Check out that box underneath – as a reference it now makes a lot more sense, eh?)

It produces sine waves and square waves up to 25 MHz, and has tons of other waveforms built in, including ramp, triangle, pulse, noise, and more. In fact, since it’s an AWG, you can load any waveform shape into it, and it’ll reproduce it at up to 125 Mega-samples per second and 14-bit resolution (it goes up to 6 MHz in this mode).

Two other major capabilities of such a unit are: the ability to “sweep” across a range of frequencies and being able to “modulate” the generated signal with another one in numerous ways: AM, FM, PM, PWM, and FSK.

As with the Hameg HMO2024 oscilloscope and the GW-Instek GPD-2303S power supply, this thing can be remotely controlled over USB. So it can be driven from a computer to perform complex and/or lengthy tests.

This model does more than I need, but there was a good “price burner” offer at Distrelec, so I decided to go for it. Function generators are not the most important instruments for an electronics lab, but they are extremely useful to learn about all sorts of analog electronics, and to illustrate various concepts and effects “for real”. Note that for lower frequencies, you can generate rough arbitrary waveforms with simply an ATmega and a few resistors.

Here’s the FFT spectrum of its 25 MHz sine wave – a few spikes at 25 MHz multiples, as expected, plus a bunch of 90..105 MHz spikes which also appear when the AWG output is off (more about those tomorrow):

Such an AWG is not limited to strictly analog uses, by the way. This unit should also be able to generate a serial bit-stream, like an RS232 message, for example. Such patterns can be loaded via USB on the front panel.

I intend to put this instrument to good use here at JeeLabs, not in the least to create good examples for future weblog posts and to illustrate relevant electronics concepts in that huge playground called Physical Computing.

A week ago, there was a post about various clock options and their accuracy.

These clocks generate a stable pulse or sinewave, basically. But what if you need a different frequency?

Suppose you get a very accurate 1 pulse-per-second (i.e. 1 Hz) signal from somewhere, but you want to keep track of time in microseconds? IOW, you need a 1 MHz clock, preferably just as accurate. One way to do this, is to use a “Voltage Controlled Oscillator” (VCO). It can be any frequency really – the idea is to divide its output down to 1 Hz and then compare it with your reference clock. If it’s either too slow or too fast, adjust the voltage used to set the precise frequency of the VCO, and bingo – within no time (heh, so to speak), your VCO will be “locked” onto the reference and generate its target frequency, at just about the same accuracy as the 1 pps reference.

My Rubidium clock came with a 63.8976 MHz VCO as part of the bargain:

With no control voltage it generates a sinewave-ish very high frequency signal from just a 3.3V power supply:

That frequency is not as awkward as it looks: 638976 = 3 * 13 * 16384, so you can get 100 Hz out of it with a few simple dividers, as well as any integral fraction of that (including 1 Hz). Another way of going about this is to divide the clock by a simple power of two, say 256 or 4096, and then pass the resulting square wave to an ATmega’s timer/counter input. I haven’t hooked up this VCO to the Rb clock yet, since there’s a bit more logic involved – look up “phase locked loop” (PLL) if you’re interested.

Another source of very stable clock signals is the GPS navigation system (see also this note). Their clocks are used to be made a little bit jittery for civilian use, but this averages out over time, so you can still lock onto it and get a very accurate long-term reference. Look up Allen variance to find out more about short- vs long-term stability – it’s fascinating stuff, but as with most things: once you get into the details it can become quite complex.

To summarize: with a VCO you can produce any frequency you like given some stable reference. So I’m happy with my 10 MHz @ 10 ppt atomic clock, for those rare cases when I’ll need it. And for its geek factor, of course…

Do all these extreme accuracies matter? Well, apart from TDMA, think of it this way: an 868 MHz RFM12B wireless radio with 1 ppm accuracy may be off by 868 Hz. That’s no big deal because the RFM12B’s receiver uses Automatic Frequency Control (AFC) to tune itself into the incoming signal, but with bandwidths in the kilohertz range, you can see that all of a sudden a couple of ppm isn’t so academic any more!

There are several scenarios where it’d be nice to detect the pulse of a blinking LED – especially low-power, because then we can sense it with a long-lasting battery-powered setup, such a JeeNode or JNµ.

Fortunately, that’s fairly easy to do. I used this test setup to try things out:

The left-hand side is a test pulse, generating 10 ms pulses once a second to simulate a typical indicator light. It’s simple enough with no further explanation needed.

The right-hand side of the above circuit is the actual pulse sensor we’re trying to implement. It’s a voltage divider with on the upper half a fixed resistor (well, a trimmer, but we only have to adjust it once) and the lower half is a Light Dependent Resistor (LDR) – like these two examples:

We want to generate one electrical pulse for each incoming light pulse, in such a way that it could trigger an ATmega’s digital input pin. With a clean pulse we could then set up a pin-change interrupt and keep the ATmega asleep most of the time.

The trouble is that LDR’s and voltage dividers are analog i.s.o digital. One way would be to constantly read out the signal as analog input. But this sort of polling and continuous ADC use eats up quite a bit of power – a digital signal would be a lot better as it’d allow us to use pin-change interrupts.

No worries. A digital signal is also a voltage, but it has to stay under a certain limit to be treated as digital “0” and above another limit to act as digital “1”. Here are the specs from the ATmega328 datasheet:

With a JeeNode running at 3.3V, we get: “0” ≤ 1V and “1” ≥ 2V. Note that in theory voltages between 1 and 2V will have indeterminate results, but in practice the signal will work fine as long as it doesn’t stay forever within that gray zone.

The trick is to make that LDR sensor as sensitive as possible. The LDR which I used is a fairly standard one (same one as included with the Room Board) and rises to over 1.5 MΩ resistance when dark. Let’s assume 1 MΩ as extra margin, then we could use 470 kΩ as upper resistor of the above resistor divider, and the resulting signal would be about 2.2V when dark.

The way I maximized the dark-state resistance was to place it in a small black plastic cap, as shown in the above photograph. This is essential, as you’ll see.

Now the actual pulse detection: the resistance of an LDR drops (quite dramatically) in the presence of light, so the trick is to place it close enough to the blinking LED that we want to “read out”. I placed my blinking test LED a few millimeteres from the black cap (which is open at the end, of course);

Here’s a scope snapshot of the LED pulse (channel 1, yellow trace) and the detected signal (channel 2, blue trace):

You can see the LDR signal dropping when light is detected, and that the LDR actually needs a bit of time to react. For 10 ms pulses, it’s plenty fast enough, though.

This configuration is probably ok – the voltage swings from about 1.8V (a marginal “1”) down to 0.7V (a clean “0”). The whole setup really depends on first getting the dark resistance as high as possible (i.e. shielded from any stray light) and pulling it down enough during the LED blink (i.e. close enough to pick up a good LED signal).

When the LED is inserted inside the plastic tube, the signal becomes much stronger – but recovery is slower:

It all hinges on the pull-up resistor, really. Which is why the best way to create this sensor is to use an adjustable 1 MΩ trimpot, and tweak it. You won’t need an oscilloscope or even a multimeter to get optimal results:

• very important: shield the LDR from stray light as well as you can
• pick as high a resistance as possible which still gives a “1” signal (between 100 kΩ and 1 MΩ)
• place the LDR + shield near enough to the LED to generate a “0” pulse
• tweak and iterate the above steps until it works reliably under all conditions

For minimal power consumption, the pull-up resistor should be as large as possible. Example: with an optimal pull-up of 1 MΩ and the LDR’s dark resistance about 1 MΩ as well, the quiescent current draw will be (Ohm’s law: I = E/R) 3.3 V / 2 MΩ = 1.65 µA, an excellent value for ultra-low power nodes. During the LED light pulse, this will increase to at most twice that (i.e. if the LDR resistance were to drop completely to 0 Ω).

Note that a more sensitive sensor design will be needed if you want to actually measure the length of the pulse with a decent accuracy, but for simple counting purposes where incident light can be kept out, there is nothing simpler than this LDR + pull-up trimmer, probably.

Update – More info about LDR’s on the LadyAda site.

Someone drew my attention to a very small PIR sensor on eBay:

The size is great, of course – but the current consumption isn’t: I measured 1.9 mA idle current @ 5V.

The other inconvenience, in the context of JeeNodes, is that this sensor expects a 5..9V supply voltage.

Using my new accurately adjustable power supply, I was able to establish that it actually works all the way down to 3.6V – with current consumption down to 1.3 mA. But that’s still far from the 50 µA current consumption of the PIR sensor used in the Room Board, so this rules out ultra-low power battery nodes. The detection range is specified as 2 to 3 m, not stellar but probably enough for many uses.

Here are the two sides of this really tiny sensor (whoops, I accidentally cracked the lens):

The chip marked 7144-1 appears to be a 4.4V LDO regulator, with excellent < 5 µA idle current and 0.06 V drop-out voltage under light load, but that seems to point to a circuit which really expects to run at 4.4V internally.

I have no idea what this spec means on the eBay page:

It’s definitely not referring to the idle power consumption of this sensor. Too bad!

Should we try and design our own PIR sensor? I wonder what that would take – some way to stabilize on an average detector level, and then detecting changes in that value? Using an ultra-low power op-amp?

My existing lab power supply delivers 30V @ 3A, which is more than enough for normal use, but it uses linear fine + coarse potentiometers, which are in fact not optimal for really fine adjustments. I’ve been using it a lot, and I really have been wanting something more convenient for quite some time.

So I decided to get a second and more high-end unit, the GW-Instek GPD-2303S:

It even comes with a “calibration certificate”, FWIW:

There are many lab power supplies out there, and I intend to come up with a really good option for low end use in the context of the Thursday Toolkit series, but I’ve got enough future projects piled up here to justify this instrument for JeeLabs. Everything other than the ultra-low power experiments will benefit from this.

BTW, if you’re looking for a DIY design which is coming along very nicely, check out the EEVblog episode list, where Dave Jones has over half a dozen fascinating videos about how he is designing a really nice Arduino-compatible power supply, with all the bells and whistles you might be after: finely programmable voltage and current range, an LCD display, rotary switches for adjustment, etc. Here’s the first one in the series.

The GPD-2303S delivers 2x 30V @ 3A, i.e. up to 180W of controlled DC power. There’s a 3-channel unit with extra 2.5/3.3/5V output, even a 4-channel unit, but I’ve got enough supplies here now to cover such needs.

Note that lab power supplies are designed to “float” w.r.t. ground. The reason for this is described in my two weblog posts here and here. So you can hook them into your setup in any way you like. Even doing some totally crazy stuff like a adding a 50V DC component to AC mains would be possible…

Anyway. The nice thing about this supply (even though its shape is a bit deep for my workspace), is that it includes two independent supplies which can be used in series (double the voltage) or in parallel (double the current) to get 0..60V or 0..6A capability, and that both voltage and current can be controlled and measured very accurately (not quite down to the 1 mV/mA levels as they claim, but close).

This power supply is not a really high-end one, though (which would cost even more), since there is no remote sensing, for example. So small losses over the cabling are not compensated for. I’m not too worried, because with large currents I’m usually not really concerned about 10 mV error.

More important is that it’s a linear power supply with only 1..2 mV ripple, and that the current limit can accurately be set to very low levels. By setting it to 50 mA, say, you can avoid most damage when hooking up things the wrong way – as so often happens while messing around with circuits.

Also very nice is that this unit is programmable – meaning that you can control it fully via USB. That opens the door to all sorts of stress and limits testing, i.e. plotting the effects of a slow voltage ramp on a circuit, for example.

Sooo… with this new addition to JeeLabs, I hope to stay out of Mr. Murphy’s path a bit more!

Tracking time is as old as… well, time itself really. FWIW, I stopped wearing a wristwatch about a year ago. When traveling, I often don’t have a convenient way to tell the time with me. I like it that way because it pushes me to leave a bit earlier and enjoy the journey a bit more, instead of stressing out to reach some location on earth at some particular point in time. “Onthaasten” as the Dutch say (“un-hurry”).

That wristwatch was one of the most beautiful time-pieces I ever owned, and darn accurate – less than a minute off per year. More than accurate enough for day-to-day use without ever adjusting it (except for DST).

Still, time is everything. Cell phones make very efficient use of bandwidth (and energy) by using time-division multiple access (TDMA), i.e. taking up specific time slots to get a transmitter to talk to the receiver it wants to reach, without collisions. It’s a common technique in many advanced networks, not just with cell phones.

TDMA requires all parties to be aware of time. No wonder that cell-phone towers need the Rubidium clock I described in the past two days. If your timing is off, you end up jamming others, and to avoid that the system then needs to introduce wider gaps around each slot. More gaps = more time & energy wasted, as each unit has to wait longer in receive mode to be certain it picks up the entire packet, and more gaps = more unused bandwidth.

For good timing, you need to have every node in sync within a millisecond or so. Perfect timing means a receiver can turn on exactly when the transmitter starts, and switch off right after the end. But the need for exact time is bad news for the simplest ultra-low power nodes, which tend to use an RC-controlled watchdog timer for the sleep modes. On an ATmega, watchdog accuracy is only about 10% in the worst case:

Ok, time to get into some terminology…

• one percent (%) is 1 per 100, of course
• one part per million (ppm) is 0.0001 percent
• one part per billion (ppb) is 0.0000001 percent
• one part per trillion (ppt) is 0.0000000001 percent

It’s easy to make mistakes with so many zero’s, so let’s approach it from another angle: a year has about 31.5 million seconds, so let’s specify time accuracy in the amount of error over a year. And let’s not fuss over 50%, I’ll round things up or down a bit for convenience.

My trusty old Seiko Lasalle wristwatch:

• estimated 1 min/year, i.e. an astoundingly good 2 ppm

ATmega watchdog accuracy:

• worst case: 10 % = can be over a month per year off
• if supply is 3.3V ± 0.1V and temp is 25°C ± 10°C: 1 % = within 3 days per year

The 16 MHz resonator used in a JeeNode:

• 0.5 %, i.e. less than 2 days per year off

The crystal normally used in RBBB’s, Arduino’s, RTC’s etc:

• 50 ppm, i.e within half an hour per year

The more accurate crystal used in a JeeLink:

• 10 ppm, i.e. within 5 minutes per year

The Precision RTC Plug, which will be released later:

• 2 ppm, i.e accurate within 1 minute per year

The calibrated Temperature Compensated Crystal Oscillator (TCXO), mentioned recently:

• 0.1 ppm, i.e. within 3 seconds per year

And then that new Rubidium clock:

• 10 ppt, off by no more than 0.3 milliseconds per year

Cesium clocks can do even better, by the way:

• 0.01 ppt, no more than 0.3 microsecond per year

And how’s this Quantum Logic Clock, mentioned in one of the recent comments:

• less than 0.3 nanoseconds off per year!

The clocks mentioned so far all try to be spot on as best as they can. That’s what a clock is for, after all.

But that’s not all there is. Next week I’ll describe how a fixed reference can be used to stabilize any frequency.

After yesterday’s intro of my “get your own atomic clock”, which is really just doodling, here’s the next step:

The clock, and the PCB panel it came attached to, has been placed in an all-plastic enclosure along with a little 15V @ 1.7A switching power supply. This thing needs quite a bit of power and actually gets quite hot. Nevertheless, I expect that placing it inside this relatively small plastic enclosure will not be a problem because much of the heat seems to be generated simply to keep the Rubidium “physics package” inside at a certain fixed temperature. For that same reason, I suspect that the heat sink on which this clock is mounted is not so much meant to draw heat away, but to maintain a stable temperature and improve stability.

Speaking of stability… here are the specs of this unit from eBay:

To get an idea: 10 to the power -11 frequency stability is less than 0.3 milliseconds per year error!

This particular unit (they are not all identical, even when called “FE-5680A”) also needs a 5V logic supply.

I haven’t yet decided how to bring out various signals, so I’ll hook up the 50Ω BNC connector on the back first and wait with the rest. Also needed: a LED power-on light, LED indicators for the “output valid” and “1 pulse-per-second” signals (via a one-shot to extend the 1 µs second pulse), and a 7805 regulator. Here’s the front – so far:

I don’t intend to keep this energy-drain running at all times, but it’ll be there at the flick of a switch to generate a stable 10 MHz signal when needed. One of the things you can do with it is calibrate other clocks, and compare their accuracy + drift over time and temperature.

Geeky stuff. For a lot more info about precise time and frequency tracking, see the Time Nuts web site.

Tomorrow, I’ll describe some of the trade-offs w.r.t. time for JeeNodes and wireless sensors.

Triggered by a video on EEVblog of a Rubidium frequency standard and its teardown, I decided to get one myself. These things are available from eBay for around €40 these days, as recalls from cellphone towers, apparently:

It’s about the size of a hard drive, it’s completely closed, and there really is nothing to it – connect power, wait a few minutes for things to stabilize, and out comes a 10 MHz signal – a perfect black box:

There’s a capacitor in the output circuit, so the resulting signal is AC-coupled and hence centered around 0V and 1.5 V peak-to-peak. There’s also a 1 pulse-per-second (PPS) output signal, with a 1 µs pulse (at first I thought it didn’t work, but the pulse is really there).

The big deal about such a Rubidium-based atomic clock is its accuracy. More on this tomorrow…

I ran out of USB BUB interface boards in the shop the other day (all my fault, not paying attention), so these last few days a slightly different version is being sent out. This is an earlier, but nearly equivalent, design by Lennart Herlaar – and as it so happens, I had a bunch of them lying around – a perfect susbtitute until the BUB’s by Modern Device are back in stock next week.

Here’s the USB FTDI Board in close-up:

As with the BUB, you need to solder on the 6-pin FTDI connector for use with JeeNodes and RBBBs:

The “settings” for this board is to pass the 5V supply voltage from the USB connector (solder jumper, already installed), and to set the logic levels jumper to work with 3.3V, as required by the JeeNode (or 5V for RBBB use). It’s not that critical, really.

Sooo… slightly different unit and shape, but does the same job as before, i.e. connecting up to USB for power, as well as serial-over-USB uploading and debugging.